Charge sharing time domain filter

ABSTRACT

An approach to time domain filtering uses a passive charge sharing approach to implement an infinite impulse response filter. Delayed samples of an input signal are stored as charges on capacitors of a first array of capacitors, and delayed samples of the output signal are stored as charges on capacitors of a second array of capacitors. Outputs are determined by passively coupling capacitors of the first and second arrays to one another, and determining the output according to a total charge on the coupled capacitors. In some examples, a gain is applied to the total charge prior to storing the output on the second array of capacitors. In some examples, a charge scaling circuit is applied to the charges stored on the arrays prior to coupling capacitors to form the output.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. Pat. No. 8,717,094, whichclaims priority to U.S. Provisional Application No. 61/493,893 titled“CHARGE SHARING IIR FILTER” filed Jun. 6, 2011, and which is acontinuation-in-part of PCT Application No. PCT/US11/48278, titled“CHARGE SHARING ANALOG COMPUTATION CIRCUITRY AND APPLICATIONS”, filed onAug. 18, 2011, and published as WO2012024507 on Feb. 23, 2012, thecontents of each is incorporated herein by reference.

This application is related to, but does not claim the benefit of thefiling date of U.S. Pat. No. 8,188,753, titled “ANALOG COMPUTATION”,issued on May 29, 2012, and U.S. patent application Ser. No. 13/482,112,filed on May 29, 2012, which are also incorporated herein by reference.

BACKGROUND

This invention relates to time domain filtering and filters that makeuse of charge sharing techniques.

Time domain filters are generally classified as either finite impulseresponse filters (FIR filters) or infinite impulse response filters (IIRfilters). FIR filters compute their output signals based solely on afinite history of values of the input signal while IIR filters computetheir output signals based on values of the input signal and previousvalues of the output signal (i.e., IIR filters are recursive).

Conventional digital time domain filters receive a digitized,discrete-time (e.g., sampled) input signal and generate a digitized,discrete-time output signal which includes an altered version of thesampled input signal. Such filters are generally implemented usingdigital hardware such as dedicated digital signal processing chips(DSPs). Various designs of such filters and their associated advantages,disadvantages, and applications are well known in the art and are notfurther discussed in this application.

Discrete time, or more generally discrete sample (e.g., spatial sample),time domain filtering has been implemented using analog signals. Forexample, a wide range of what are often referred to as “switchedcapacitor” filters are used, generally making use of a technique ofcharge transfer using active amplifier stages, whereby a signalrepresented by charge on capacitive elements at an input of an amplifierstage is transferred to charge on capacitive elements at an output ofthe amplifier stage. An advantage of circuitry that directly processesanalog signals is avoiding the need to convert the signal levels todigital form and reduced circuit resources required to process thesignal levels in analog form and/or higher clocking rates, as comparedto use of a digital arithmetic unit of digital signal processor.

Another approach to discrete time analog signal processing makes use ofactive elements for combining analog signals. For example, one approachto implementing a finite impulse response filter is to use a capacitorarray (e.g., a tapped delay line) to store signal values, and a set ofanalog multipliers with controllable gain that scale the voltages at theoutputs of the array, and/or integrators prior to combination todetermine the output of the filter.

SUMMARY

In one aspect, in general, an approach to time domain filtering uses apassive charge sharing approach to implement an infinite impulseresponse filter. Delayed samples of an input signal are stored ascharges on capacitors of a first array of capacitors, and delayedsamples of the output signal are stored as charges on capacitors of asecond array of capacitors. Outputs are determined by passively couplingcapacitors of the first and second arrays to one another, anddetermining the output according to a total charge on the coupledcapacitors. In some examples, a gain is applied to the total chargeprior to storing the output on the second array of capacitors. In someexamples, a charge scaling circuit is applied to the charges stored onthe arrays prior to coupling capacitors to form the output.

In another aspect, in general, a signal processing device has a firstdiscrete time analog signal filter section. This first section includesan input for accepting a time series of input signal values, an outputfor providing a time series of output signal values, an analog signalstorage section comprising a plurality of capacitors, and a number ofswitching elements. The switching circuit elements are configurable to(a) charge successive subsets of capacitors of a first plurality ofsubsets of the plurality of capacitors according to successive values ofthe input signal values, (b) couple successive subsets of two or morecapacitors of a second plurality of subsets of the plurality ofcapacitors to form successive values of a time series of intermediatesignal values, and (c) charge successive subsets of two or morecapacitors of a third plurality of subsets of the plurality ofcapacitors according to successive values of the intermediate signalvalues. The section also includes circuitry for forming the time seriesof output signal values according to the time series of intermediatevalues, and control logic for controlling configuration of the switchingcircuit elements in successive phases of a clock signal to form the timeseries of output signal values as an application of a desired infiniteimpulse response filter to the time series of input signal values.

Aspects can include one or more of the following features.

The signal storage section comprises a first storage section and asecond storage section, wherein (a) the subsets of capacitors of thefirst plurality of subsets are formed from capacitors of the firststorage section, (b) each of the subsets of the second plurality ofsubsets is formed from capacitors in both the first storage section andthe second storage section, and (c) the subsets of the third pluralityof subsets are formed from capacitors of the second storage section, and

The device further includes active circuitry for charging the successivesubsets of the third plurality of subsets of capacitors according to thesuccessive intermediate signal values.

The switching circuit elements are configurable to cause at least someof the charges on the capacitors in the storage section to be scaledaccording to configurable factors.

The device further includes at least a second discrete time analogsignal filter section coupled to the first filter section such that thetime series of output values of the first section form a time series ofinput values for the second section.

The first filter section and the second filter section each areconfigurable to implement infinite impulse response filters.

The device further includes at least a second discrete time analogsignal filter section coupled to the first filter section such that atime series of output values of the second section forms the time seriesof input values for the first section.

The second filter section includes an FIR decimation filter.

The first filter section implements an infinite impulse response filterwith delay terms limited to a delay of samples, and the first storagesection includes an array of capacitors and the second storage sectionincludes an array of capacitors.

The device further includes an input for a clock signal, and wherein thecontrol logic comprises digital storage for configuration data, andlogic circuitry for combining the clock signal and the configurationdata to control the switch circuit elements.

Aspects may include one or more of the following advantages.

Passive scaling circuits based on charge sharing techniques can be usedto implement time and frequency domain digital filter designs. Suchimplementations can provide low power and filtering with a smallfootprint in signal processing applications such as hearing aids or thefront ends of analog to digital converters (ADCs).

As example applications, filters using such charge sharing can be usedas anti-alias filters to prevent aliasing in acquired signals, notchfilters which can remove unwanted signal components such as linefrequency hum (e.g., 60 Hz hum). In other examples, high-pass filtersusing charge sharing techniques can be used to eliminate baseline wander(i.e., DC offset) in signals before they are digitized.

In some examples, the approaches described above can be used toimplement a configurable digital filter design on a self-containeddevice such as an integrated circuit. Use of such an integrated circuitcould allow system designers to save cost and limit power consumption byavoiding the need to use digital signal processing hardware.

Other features and advantages of the invention are apparent from thefollowing description, and from the claims.

DESCRIPTION OF DRAWINGS

FIG. 1 is a time domain filter system.

FIG. 2 is a fixed second order time domain IIR filter.

FIG. 3 is a configurable second order time domain IIR filter.

FIG. 4 is a detailed view of a configurable second order time domain IIRfilter.

FIG. 5 illustrates a memory loading phase of a configurable second ordertime domain IIR filter.

FIG. 6 illustrates a first charge transfer phase of a configurablesecond order time domain IIR filter.

FIG. 7 illustrates a second charge transfer phase of a configurablesecond order time domain IIR filter.

FIG. 8 illustrates an output phase of a configurable second order timedomain IIR filter.

FIGS. 9 a-9 d illustrate a first mode for applying a single filtercoefficient of the configurable second order time domain IIR filter.

FIGS. 10 a-10 d illustrate a second mode for applying a single filtercoefficient of the configurable second order time domain IIR filter.

DESCRIPTION 1 System Overview

Referring to FIG. 1, an exemplary a filter system 100 receives an analoginput signal 102 and a clock signal 101 as inputs, and applies adiscrete time filter to the input signal to provide an output signal104. The filter system 100 implements a charge sharing discrete timeanalog filter which processes the input signal 102 according to a filterconfiguration instruction/data 107 provided from an external filterconfiguration module 106. In this example, the output signal 104 ispassed to a downstream component, in this case, an analog to digitalconverter 118. It should be understood that the filter system 100 ismerely one example of a configuration of such a filter provided toillustrate the techniques used, and it should be understood that otherexamples have different arrangements of these and/or other modules.

The exemplary filter system 100 is assumed to receive a band-limitedcontinuous time signal, for example, having been filtered by aconventional continuous time analog filter prior to being received bythe filter system. In this example, the first stage of the system is acharge sharing discrete time decimation filter 110 which samples theinput signal 102 at a first sampling rate, and provides discrete timeanalog values at a second lower sampling rate, for example, with an 8×decimation rate. The decimation filter 110 is followed by a first chargesharing discrete time second order IIR filter 112 cascaded with a secondcharge sharing discrete time second order IIR filter 114, each of whichfilters the signal at rate of the output of the decimation filter. Theoutput of the second IIR filter is then passed to a charge sharingdiscrete time interpolation filter 116, which outputs an discrete timeinterpolation of the signal at a third sampling rate (e.g., at the samesampling rate as the input is sampled).

As introduced above, the decimation filter 110 reduces the sampling rateof the input signal 102 by an integer factor (e.g., by a factor of 8).The decimation filter 110 first samples the input signal 102 at a samplerate which is dictated by the clock 101 by charging a different set ofsample capacitors (not shown) for each cycle of the clock 101. Thedecimation filter 110 then generates a decimated signal 111 which has adecimated sampling rate that differs from the clock frequency by aninteger value. To avoid aliasing, the decimation filter 110 also appliesa low-pass filter to the anti-aliased signal 108 to remove anyfrequencies which are greater than ½ of the decimated sampling rate.

In some examples, the charge sharing discrete time decimation filter 110is implemented as a charge sharing discrete time FIR filter as isdescribed in U.S. Pat. No. 8,188,753 titled “Analog Computation” whichis incorporated herein by reference. One approach described in theprevious patent is an N-tap FIR filter implemented using N² capacitors,whose values are chosen (e.g., fabricated to be fixed) to match thecoefficients of a desired filter. For example, in the case of thedecimation filter for the present example, an N=128 tap lowpass windowfilter may be used. Note that in the technique described in the previouspatent, each input sample is used to charge N of the N² capacitors, andeach output is determined by charge sharing a different N capacitors. Inthis case with decimation by a factor K (e.g., K=8), although eachoutput is determined by sharing charges on N capacitors, because onlyone in K outputs are required, each input only has to charge N/Kcapacitors, and only N²/K total capacitors are needed to store inputvalues before combination through sharing.

In general, various examples of the filter systems includeinfinite-impulse-response modules. These modules can have fixedcharacteristics, or can be configurable prior to operation. In theexemplary filter system 100 shown in FIG. 1, the decimated signal 111 ispassed to the first charge sharing discrete time analog IIR filter 112(referred to as the ‘first biquad’ 112). As is well known in the art, asecond order IIR filter can be represented in the Z-transform domain bythe equation:

${h(z)} = \frac{b_{0} + {b_{1}z^{- 1}} + {b_{2}z^{- 2}}}{1 + {a_{0}z^{- 1}} + {a_{1}z^{- 2}}}$

with the filter output, y[n], represented in the discrete time domain bythe equation:

y[n]=b ₀ x[n]+b ₁ x[n−1]+b ₂ x[n−2]−a ₁ y[n−1]−a ₂ y[n−2]

Before the filtering operation of the system 100 the desired values ofthe coefficients b₀, b₁, b₂, a₁, and a₂ are processed by the filterconfiguration module 106 to determine the configuration instruction/data107, which are passed to the system 100. During the filtering operation,the first IIR filter 112 generates a first filtered signal 113 byfiltering the decimated signal 111 using charge sharing discrete timeanalog filtering techniques, which are described in detail below.

The first filtered signal 113 is passed to the second charge sharingdiscrete time analog filter, which in this example is also an IIR filter114 (referred to as the ‘second IIR filter’ 114) which generates asecond filtered signal 115. The second IIR filter 114 has the same basicstructure as the first IIR filter 112. The coefficients of the secondsecond order IIR filter 114 (i.e., b₀, b₁, b₂, a₁, and a₂) are assignedseparately from the coefficients of the first IIR filter 112 by thefilter configuration module 106. By cascading the two second order IIRfilters 112, 114 as is shown in FIG. 1 a fourth order IIR filter can beimplemented. As introduced above, in other examples, even more filtersare cascaded and/or otherwise interconnected to implement other types offilters.

The second filtered signal 115 is passed to the charge sharing discretetime interpolation filter 116 which generates the output signal 104 byup-sampling (e.g., by a factor of 8) by interpolating the secondfiltered signal 115. As with the decimation filter, the interpolationfilter 116 can implemented as a charge sharing discrete time FIR filteras is described in U.S. Pat. No. 8,188,753 titled “Analog Computation”.One approach described in this patent is a N-tap FIR filter implementedusing N² capacitors, whose values are chosen (e.g., fabricated to befixed) to match the coefficients of a desired filter. In some examples,each output of the second IIR filter is replicated K times, beforefiltering, while in other examples, each output of the IIR filter ispadded with K−1 zero values before the next output of the IIR filter. Inone example of such an interpolation filter, a windowed ideal FIR filteris used. In another example, linear interpolation of the samples at Ktimes the rate of the IIR filters can be implemented with an N=2K pointFIR filter. Each output sample of IIR filter then charges N capacitors,and each interpolated output sample is formed by charge sharing of 2capacitors, thereby making use of an array of 2K by 2 capacitors. Otherforms of interpolation filtering, which do not necessarily make use ofcharge sharing can also be used. In some examples, the interpolationfilter 116 is adaptive to match the characteristics of the outputsignal.

In some examples, the exemplary filter system 100 described above isimplemented as a separate package which can be included as a module inlarger systems. For example, the filter system can be implemented as anintegrated circuit which is packaged as a dual in line package (DIP).

2 Infinite Impulse Response (IIR) Filters

Having described the exemplary filter system 100 above, we now describethe general approach to forming infinite impulse response filters, ofwhich the IIR filters 112 and 114 of the system 100 are examples.

2.1 Fixed IIR Filter Architecture

Referring to FIG. 2, a fixed IIR filter 212 (illustrated here as asecond order IIR filter (N=2) structure, as in FIG. 1, understandingthat this is just an example of possible numerator and denominatorpolynomial degrees) receives an input signal, x[n] 211 and filters theinput signal, x[n] 211 according to a predetermined, fixed transferfunction

${h(z)} = \frac{b_{0} + {b_{1}z^{- 1}} + {b_{2}z^{- 2}}}{1 + {a_{0}z^{- 1}} + {a_{1}z^{- 2}}}$

to generate a filtered output signal, y[n] 213. The IIR filter 212 is asubstantially passive circuit in that it includes few (i.e., one) activegain element in the signal path. The IIR filter 212 includes a passivenumerator processor 1022, an amplifier 1038, and a passive denominatorprocessor 1024, which are described below.

The input signal, x[n] 211 is first passed to the numerator processor1022. The numerator processor 1022 includes a first analog memory 1030,which for a numerator degree N has (N+1)² capacitors. In this fixedfilter structure, the capacitors are chosen according to thecoefficients b₀ to b_(N). For example, the capacitors are indexed from(0,0) to (N, N), and the input time n charges capacitors (k, (n−k) mod(N+1)), for k=0, . . . , N, illustrated by the diagonal line 1074through the analog memory in FIG. 2.

In determining the output of the numerator processor 1022 for time n,N+1 capacitors of the numerator processor, with index values (k, n mod(N+1)), for k=0, . . . , N (forming a column) are coupled via a sharingnode 1034, which permits bidirectional flow of charge between its portsas voltage on the coupled capacitors equilibrates. In this fixedstructure, the capacitors are chosen such that the size of thecapacitors with index (k, *) have values proportional to b_(k). Due tothe differing capacitances of the capacitors 1076 in the first analogmemory 1030, a different amount of charge is generally stored on eachcapacitor 1076 included in the diagonal line 1074. Thus, the chargestored on each capacitor 1076 can be seen as a weighted input sample ofthe input signal x[n]. The charges of the capacitors 1076 included in acolumn of capacitors 1076 of the first analog memory 1030 represent aweighted time series of the input signal (i.e., b₀x[n], b₁x[n−1], andb₂x[n−2]). The charges on the column of capacitors 1076 are coupled tothe sharing node 1034 which essentially acts as presenting a sharedtotal charge proportional to b₀x[n]+b₁x[n−1]+b₂x[n−2].

The numerator output signal 1036 is passed to an output sharing node1037 along with a denominator output signal 1039, which as describedfurther below essentially presents a charge proportional to−a₁y[n−1]−a₂y [n−2]. The output sharing node 1037 combines the numeratoroutput signal 1036 and the denominator output signal 1039 by essentiallyfurther sharing charges and passes the result to an amplifier 1038. Notethat the voltage provided to the amplifier is proportional to thedesired

y[n]=b ₀ x[n]+b ₁ x[n−1]+b ₂ x[n−2]−a ₁ y[n−1]−a ₂ y[n−2]

The amplifier scales the voltage by a predetermined gain factor,resulting in a filtered output signal, y[n] 213.

Note that the sharing nodes described above are not necessarily explicitin the circuit layout of the IIR filter 212 and can equivalently bereplaced by a bus structure (i.e., two wires/traces for a differentialsignal implementation).

The filtered output signal, y[n] 213 is also fed back as an input to thedenominator processor 1024. The denominator processor 1024 implementsthe denominator portion of the filter equation shown above (i.e.,−a₁y[n−1]−a₂y [n−2]), resulting in the denominator output 1039. Thedenominator processor 1024 includes a second analog memory 1040 (of sizeN by N) and a sharing node 1048.

The second analog memory 1040 receives the filtered output signal, y[n]213 as input and stores a time series of samples of the filtered outputsignal, y[n] 213. In particular, the filtered output signal, y[n] 213 isstored in a diagonal line 1080 of capacitors 1082 in the second analogmemory 1040. For example, the capacitors are indexed from (0,0) to (N−1,N−1), and the output y[n] charges capacitors (k, (n+1−k) mod N), fork=0, . . . , N−1, illustrated by the diagonal line through the analogmemory in FIG. 2.

In this fixed structure, the capacitors are chosen such that the size ofthe capacitors with index (k,*) have values proportional to a_(k+1). Dueto the differing capacitances of the capacitors 1082 in the secondanalog memory 1040, a different amount of charge may be stored on eachcapacitor 1082 included in the diagonal line 1080. Thus, the chargestored on each capacitor 1082 can be seen as a weighted input sample ofthe filtered output signal, y[n] 213. The charges of the capacitors 1082included in a column of capacitors 1082 of the second analog memory 1040represent a weighted time series of the filtered output signal (i.e.,a₁y[n−1], and a₂ y[n−2]). The charges on the column of capacitors 1082are passed to the sharing node 1048 where they are combined to generatea denominator output signal 1039 which essentially represents a chargeproportional to −a₁y[n−1]−a₂y[n−2].

As is described above, the denominator output 1039 is passed to theoutput sharing node 1037 along with the numerator output 1036. Byselecting the gain of the amplifier 1038 and the proportionalityconstants relating the filter coefficients to the capacitor sizesappropriately, the output of the sharing node 1037 is a voltageproportional to the desired output

y[n]=b ₀ x[n]+b ₁ x[n−1]+b ₂ x[n−2]−a ₁ y[n−1]−a ₂ y[n−2].

Note that the arrangement of capacitors can be modified, essentially bypermuting the locations of the capacitors is a rectangular array, forexample, for that the input charges capacitors in one row, and output isdetermined by capacitors in one column. However, by arrangement of thecapacitor values, the same functionality can be obtained.

There are alternative approaches to providing the voltage gain ofamplifier 1038. One approach is to amplifying the total charge sharingoutput is to use a charge transfer approach in which the total charge onthese shared capacitors is transferred to a capacitor with a capacitancesmaller than the net capacitance of the shared capacitors, therebyproviding a voltage gain.

3 Configurable IIR Architecture

One approach to providing a configurable IIR filter architecture, whichwill not be discussed further, is to simply make use of memory arrays ofconfigurable capacitors in the analog memories shown in FIG. 2. Forexample, each capacitor may include set (e.g., 8) of capacitors thatscale as factors of 2, thereby providing a discrete set of possiblecoefficient values (e.g., 256 different coefficient values). In someimplementations, the degree of configurability may not be sufficient orthe range of capacitor sizes required to be fabricated may result inundesirable characteristics (e.g., circuit size, noise, etc.).

A configurable approach for IIR filters described in detail below makesuse of analog memories with fixed and uniform capacitors, for example,with all (N+1)² and N² capacitors having the same value. Generally,prior to sharing charges from the numerator and denominator analogmemories, modified charges are formed using a multiple phase chargescaling circuit approach shown in PCT Application No. PCT/US11/48278,titled “CHARGE SHARING ANALOG COMPUTATION CIRCUITRY AND APPLICATIONS”,thereby providing a configurable scaling of the charges stored in theanalog memories to implement desired IIR filter transfer functions.

Referring to FIG. 3, a configurable IIR filter, illustrated for the N=2degree as a second order IIR filter 212 receives an input signal, x[n]211 and a configuration instruction 207 as inputs. The IIR filter 212 isconfigured as described below by the configuration instructions/data 206to implement a specified transfer function. After the IIR filter 212 isconfigured, the IIR filter 212 generates a filtered output signal, y[n]213. The IIR filter 212 includes a digital control module 220, anumerator processor 222, an amplifier 238, and a denominator processor224.

The digital control module 220 receives the configurationinstructions/data 207 and uses it to generate a numerator configurationinstructions/data 226 for the numerator processor 222, a denominatorconfiguration instructions/data 228 for the denominator processor 224and an amplifier configuration instruction 229 for the amplifier 238.

The numerator processor 222 includes a first analog memory 230 of (N+1)²equal capacitors, and (N+1) charge scaling circuits 232, 233, 235 and acharge sharing node 234. The charge scaling circuit is a configurablepassive scaling circuit that provides hybrid behavior using differentsequences of charge sharing phases, implemented using switches thatcouple successive sets of capacitors in the circuit.

During a filtering operation, the first analog memory 230 receives theinput signal 211 and stores a time series of samples of the inputsignal, x[n] 211. As is described in more detail below, the analogmemory 230 includes a number of capacitors in which the time series ofthe input signal 211 is stored. In the IIR filter 212 of FIG. 3, thetime series stored by the first analog memory 230 of the numeratorprocessor 222 in a manner similar to that for the fixed filter shown inFIG. 2, with the input at time n charging capacitors with indices (k,(n−k) mod (N+1)), for k=0, . . . , N. Note that the charges on thesecapacitors after charging are proportional to the input voltage, but areindependent of the filter coefficients at this point. Prior to couplingcharges via the sharing node 234, scaled charges are determined from thecharges on the capacitors with indices (k, n mod (N+1)), for k=0, . . ., N according to the desired filter coefficients b₀ to b_(N).

As an example, the first charge scaling circuit 235 is configured toscale its input charge by the coefficient b₀. As is described in detailbelow, this scaling operation is performed using a sequence of chargesharing phases, resulting in a charge that is proportional to originalcharge on the memory capacitor times the coefficient b₀ being present onone or more capacitors that are then coupled to the sharing node 234.The scaled outputs of the charge scaling circuits are provided to thesharing node 234 which effectively combines the scaled charges,resulting in a numerator output 236, which effectively acts as a chargeoutput proportional to b₀x[n]+b₁x[n−1]+b₂x[n−2].

The numerator output 236 is passed an output sharing node 237 where itis combined with a denominator output 239. The result of the outputsharing node 237 is passed to the amplifier 238 where it is scaledaccording to the amplifier configuration instruction 229, generating thefiltered output signal, y[n] 213.

As in the fixed IIR filter described above, the filtered output signal,y[n] 213 is passed out of the IIR filter 212 and is also fed back as aninput to the denominator processor 224. The denominator processor 224uses the inputs to implement the denominator portion of the IIR filterequation shown above (i.e., −a₁y[n−1]−a₂y[n−2]), resulting in thedenominator output 239. The denominator processor 224 includes a secondanalog memory 240, N charge scaling circuits 244, 246, and a sharingnode 248.

The second analog memory 240 has N² fixed capacitors, receives thefiltered output signal, y[n] 213 as input, and stores a time series ofsamples of the filtered output signal, y[n] 213. In the IIR filter 212of FIG. 3, the time series stored by the second analog memory 240 of thedenominator processor 224 includes the present value of the filteredoutput signal 213, y[n], along with two previous values of the filteredoutput signal 213, y[n−1] and y[n−2]. y[n−1] and y[n−2] are each passedto a corresponding one of the charge scaling circuits 244, 246 alongwith the denominator configuration instruction 228. Based on thedenominator configuration instruction 228, each charge scaling circuit244, 246 is configured such that its received sample is scaled by aspecific filter coefficient. In this example, the fourth charge scalingcircuit 244 is configured to scale its input sample by the coefficienta₁ and the fifth charge scaling circuit 246 is configured to scale itsinput sample by the coefficient a₂. As was the case above, this scalingoperation is performed using a sequence of charge sharing phases. Thescaled outputs of the charge scaling circuits 244, 246 are provided tothe sharing node 248 which combines the two outputs resulting in thedenominator output 239. The denominator output 239 is equal to thedenominator of the IIR filter equation, −a₁y[n−1]−a₂y[n−2].

As is described above, the denominator output 239 is passed to theoutput sharing node 237 along with the numerator output 236. The resultof the output sharing node 237 is scaled by the amplifier 238. Sincecharge sharing alone can only implement a limited set of filtercoefficients, the amplifier 238 is used to provide a charge buffer orgain. For example, due to the nature of charge sharing, a coefficient ofb₀≧1 is not possible without the use of an amplifier 238. The resultingoutput signal is

y[n]=b ₀ x[n]+b ₁ x[n−1]+b ₂ x[n−2]−a ₁ y[n−1]−a ₂ y[n−2].

4 Detailed IIR Filter Architecture

Referring to FIG. 4, a detailed architecture of the IIR filter 212 ofFIG. 3 is illustrated. For the sake of simplicity, the digitalcontroller 220 and its associated control signals are omitted from FIG.4. Dashed lines are used to illustrate which portions of thearchitecture of FIG. 4 correspond specific modules shown in FIG. 3. Inparticular, the numerator processor 222 is shown enclosed by a group ofconnected boxes including a box enclosing the first analog memory 230, abox enclosing the first charge scaling circuit 235, a box enclosing thesecond charge scaling circuit 233, and a box enclosing the third chargescaling circuit 232. The amplifier 238 is shown enclosed by a separatebox of dashed lines. The denominator processor 224 is shown enclosed byyet another group of connected boxes including a box enclosing thesecond analog memory 240, a box enclosing the fourth charge scalingcircuit 244, and a box enclosing the fifth charge scaling circuit 246.

Each of the analog memories 230, 240 includes a number of fixedcapacitors 350 which can be placed into various configurations using anumber of switches 352. In some examples, the analog memories 230, 240are square arrays of fixed capacitors 350 (i.e., including (N+1)²capacitors). Each of the charge scaling circuits 235, 233, 232, 244, 246includes a number (e.g., 2) of configurable capacitors 351 which arecoupled to the capacitors 350 of the analog memories and can be placedinto various configurations using a number of switches 352. Theconfigurable capacitors 351 are configurable to represent a number ofdifferent capacitance values. For example, a configurable capacitor 351may actually include six capacitors with capacitance values differing bya power of two, each capable of switching into or out of a parallelcombination with the others. In such an example, a six bit configurationword can be used to specify the capacitance value of the configurablecapacitor.

In some examples, the amplifier 238 is a differential amplifier whichoutputs both a positive version of the difference between its two inputsand a negative version of the difference between its two inputs.

In general, the IIR filter 212 computes the filtered output signal, y[n]213, in four separate phases: an analog memory loading phase, a firstcharge scaling phase, a second charge scaling phase, and a read phase.In some examples, the IIR filter 212 includes a configuration memory(not shown) which stores configuration instructions/data 107 (e.g., inflash or volatile digital memory) from the filter configuration module106. The configuration instructions/data 107 are used to configure theswitches of the IIR filter 212, causing it to cycle through the fourphases. Furthermore, the configuration instructions/data 107 may includeconfiguration words for configuring the capacitors of the charge scalingcircuits. For example, as the charge scaling circuits cycle through anumber of charge scaling phases, the configuration words can be readfrom the flash memory and used to configure the capacitance of thecapacitors.

In some examples, logic circuitry (not shown) is included in the IIRfilter 212 for the purpose of configuring the switches and capacitorsaccording to the configuration instructions/data stored in theconfiguration memory.

Each phase is briefly described below and then a detailed example ispresented.

4.1 Load Analog Memory Phase

Referring to FIG. 5, one example of loading the analog memories 230, 240is illustrated. In this example, the first analog memory 230 is a (N+1)²array of capacitors and the second analog memory 240 is a N² array ofcapacitors.

To load the first analog memory 230, a diagonal line of capacitors 454(i.e., a line crossing through C₁₃, C₂₂, and C₃₁) of the first analogmemory 230 is charged by closing switches such that the terminals of thecapacitors are electrically connected to the X+ and X− signal lines.This causes application of the input voltage to the terminals of thecapacitors which in turn places a charge on the capacitors. Since C₁₃,C₂₂, and C₃₁ all have the same capacitance, the same charge is loadedonto C₁₃, C₂₂, and C₃₁. In subsequent load phases the diagonal line ofcapacitors 454 to be loaded shifts, causing a different set ofcapacitors to be charged. When the diagonal line of capacitors 454reaches the end of the first analog memory 230, the line wraps backaround to the beginning of the first analog memory 230. In this way, atime series of the input signal, X, is stored in the first analog memory230.

To load the second analog memory 240, a second diagonal line ofcapacitors 456 (i.e., a line crossing through C₄₁ and C₄₂) of the secondanalog memory 240 is charged by closing switches such that the terminalsof the capacitors are electrically connected to the Y+ and Y− signallines. This causes application of the output voltage to the terminals ofthe capacitors which in turn places a charge on the capacitors. SinceC₄₁ and C₄₂ each have the same capacitance, the same charge is loadedonto C₄₁ and C₄₂. In subsequent load phases the diagonal line ofcapacitors 456 to be loaded shifts, causing a different set ofcapacitors to be charged. When the diagonal line of capacitors 456reaches the end of the second analog memory 240, the line wraps backaround to the beginning of the second analog memory 240. In this way, atime series of the output signal, Y, is stored in the second analogmemory 240. Note that multiple capacitors are charged with the samesample value due to the destructive nature of reading charge fromcapacitors for charge sharing phases.

Note that in this example, the X and Y signals are representeddifferentially, Thus, depending on which switches are closed, positiveor negative charges can be placed on the capacitors of the analogmemories 230, 240, according to the sign of the correspondingcoefficients b_(k) and a_(k). For example, if b_(k)<0, then capacitorsin the k^(th) row are charged with inverted inputs.

4.2 First Charge Sharing Phase

Note that prior to the charge sharing phases, the capacitors of thecharge sharing circuits are discharged using switches (not shown).

Referring to FIG. 6, after the analog memories 230, 240 are loaded, theswitches of the numerator and denominator processors 222, 224 arereconfigured to share charge between the capacitors of one column 558,560 (each column representing a time series of the input or outputsignal) of each of the analog memories 230, 240 with a first capacitorC₁₄, C₂₄, C₃₄, C₄₄, C₅₄ of each of the charge scaling circuits 235, 233,232, 244, 246.

For example, in the numerator processor 222, C₁₂ of the first analogmemory 230 is placed in parallel with C₁₄ of the first charge scalingcircuit 235, C₂₂ of the first analog memory 230 is placed in parallelwith C₂₄ of the second charge scaling circuit 233, and C₃₂ of the firstanalog memory 230 is placed in parallel with C₃₄ of the third chargescaling circuit 232.

In the denominator processor 224, C₄₂ of the second analog memory 240 isplaced in parallel with C₄₄ of the fourth charge scaling circuit 244 andC₅₂ of the second analog memory 240 is placed in parallel with C₅₄ ofthe fifth charge scaling circuit 246

Placing the capacitors in parallel as is shown in the figure causes thecharge on the capacitors of the analog memories to be distributed (i.e.,shared) between the capacitors of the analog memories and the capacitorsof the charge scaling circuits. As is described in more detail below,the amount of charge which is transferred from the capacitors in theanalog memories to the capacitors in the corresponding charge scalingcircuits depends on the respective sizes of the capacitors.

4.3 Second Charge Sharing Phase

Referring to FIG. 7, following the first charge sharing phase, theswitches of the numerator and denominator processors 222, 224 arereconfigured to implement a second charge sharing phase. Two modes ofthe second charge sharing phase are possible and each charge sharingcircuit can be configured to use a different mode.

In the first mode, switches of the numerator and denominator processors222, 224 are reconfigured to share charge between the first capacitor(i.e., C₁₄, C₂₄, C₃₄, C₄₄, or C₅₄) of a charge scaling circuit 235, 233,232, 244, 246 and a corresponding second capacitor (i.e., C₁₅, C₂₅, C₃₅,C₄₅, or C₅₅) of the charge scaling circuit 235, 233, 232, 244, 246. Inthe example of FIG. 7, the numerator processor 222 performs the firstmode of the second charge sharing phase by configuring its switches suchthat C₁₄ and C₁₅ are placed in parallel with each other and byconfiguring its switches such that C₃₄ and C₃₅ are placed in parallelwith each other.

In the second mode of the second charge sharing phase, switches of thenumerator and denominator processors 222, 224 are reconfigured to sharecharge between a capacitor of the analog memories 230, 240 (e.g., C₁₂,C₂₂, C₃₂, C₄₂, or C₅₂), the corresponding first capacitor (i.e., C₁₄,C₂₄, C₃₄, C₄₄, or C₅₄) of the charge scaling circuits 235, 233, 232,244, 246 and a corresponding second capacitor (i.e., C₁₅, C₂₅, C₃₅, C₄₅,C₅₅) of charge scaling circuits 235, 233, 232, 244, 246.

In the example of FIG. 7, the numerator processor 222 uses the secondmode of charge sharing to place C₂₂ of the first analog memory 230 inparallel with C₂₄ and C₂₅ of the second charge scaling circuit 233. Thedenominator processor 224 uses the second mode of charge sharing toplace C₄₂ of the second analog memory 240 in parallel with C₄₄ and C₄₅of the fourth charge scaling circuit 244 and to place C₅₂ of the secondanalog memory 240 in parallel with C₅₄ and C₅₅ of the fifth chargescaling circuit 246.

As was the case above, placing the capacitors in parallel as is shown inthe figure causes the charge on the capacitors distributed between eachother. As is described in more detail below, the amount of charge whichis transferred from one capacitor to another depends on the respectivesizes of the capacitors.

4.4 Read Phase

Note that during the charge sharing phases, switches s₁ and s₂ dischargethe capacitors of the amplifier 238.

Referring to FIG. 8, following the second charge sharing phase, theswitches of the numerator and denominator processors 222, 224 arereconfigured to read the scaled charges from each of the charge scalingcircuits 235, 233, 232, 244, 246.

For each charge scaling circuit 235, 233, 232, 244, 246 the switches areconfigured to implement one of two read modes. The read mode isdetermined based on the mode of the second charge sharing phase for thecharge scaling element. For example, if a particular charge scalingelement previously executed the first mode of the second charge sharingphase, then the charge on the parallel combination of the firstcapacitor (i.e., C₁₄, C₂₄, C₃₄, C₄₄, C₅₄) of the charge scaling elementand the second capacitor (i.e., C₁₅, C₂₅, C₃₅, C₄₅, C₅₅) of the chargescaling element is read as the scaled charge. If a particular chargescaling element previously executed the second mode of second chargesharing phase, then the charge on the parallel combination of acapacitor in one of the analog memories 230, 240 (e.g., C₁₂, C₂₂, C₃₂,C₄₂, C₅₂), the first capacitor (i.e., C₁₄, C₂₄, C₃₄, C₄₄, C₅₄) of thecharge scaling element, and the second capacitor (i.e., C₁₅, C₂₅, C₃₅,C₄₅, C₅₅) of the charge scaling element is read as the scaled charge.

The scaled charges are read from the charge scaling circuits 235, 233,232, 244, 245. The sum of the read charges is passed to the differentialinputs of the amplifier 238.

During the read phase, the amplifier 238, by driving its differentialinput to zero, causes the total charge on the capacitors of thenumerator and denominator processors coupled to its inputs to betransferred to the capacitors C_(A1), C_(A2) coupling its inputs andoutputs. After the read phase, the input switches s₃, s₄ of theamplifier 238 are opened (i.e., in the next load phase) and thedifferential output of the amplifier 238 is a voltage proportional toy[n]. Note that the amplifier capacitors C_(A1), C_(A2) are themselvesconfigurable, thereby controlling the magnitude of the gain of theamplifier 238.

4.5 Charge Scaling Circuit Examples 4.5.1 First Charge Scaling Circuit

Referring to FIGS. 9 a-9 d, an example of using the first charge scalingcircuit 235 of FIG. 3 to scale an input voltage of 2.0V by a filtercoefficient b₀=0.49215 is illustrated. Note that in FIG. 3, the chargesharing circuits are shown with two configurable capacitors each. InFIGS. 9 a-9 d, one capacitor configurable by setting a capacitancethrough selection of a set of (e.g., up to 6) capacitors that aremultiples of powers of two of a base capacitance. The other capacitorsimilarly has a configurable capacitor, with the addition of anoptionally configurable series capacitor (C_(c)). In the examplepresented below, the configured capacitances of the capacitors shown inthe figure are C₁₃=3.0 pF, C₁₄=2.8 pF, C_(C)=0.5 pF, and C₁₅=0.2 pF.C_(C) is a capacitor which can be configurably switched (i.e., accordingto the configuration instructions/data 207) into or out of the chargescaling circuit 235 for the purpose of providing additional scalingfactors. The serial combination of C_(C) and C₁₅ has an equivalentcapacitance of 0.143 pF.

Before the capacitor of the analog memory (i.e., C₁₃) is loaded, thecharges and voltages on all of the capacitors are assumed to be zero asis summarized in the following table:

C₁₃ C₁₄ C₁₅ + C_(C) Charge (pico-Coulombs) 0 0 0 Voltage (Volts) 0 0 0

Referring to FIG. 9 a, the capacitor in the first analog memory 230 ischarged in the memory loading phase. In this phase a first switch 862and a second switch 864 of the first analog memory 230 are closed,causing the input voltage of 2.0V to be applied to C₁₃. Applying 2.0V toC₁₃ results in a charge of 6.0 pC on C₁₃. Thus the summary of thecharges and voltages on the capacitors is as follows:

C₁₃ C₁₄ C₁₅ + C_(C) Charge (pico-Coulombs) 6.0 0 0 Voltage (Volts) 2.0 00

Referring to FIG. 9 b, the first charge sharing phase includes openingthe first and second switches 862, 864 and closing a third switch 865and a fourth switch 866, causing charge sharing between C₁₃ and C₁₄.Upon completion of the first charge sharing phase, 2.8966 pC of chargeare transferred from C₁₃ to C₁₄. The summary of charges and voltages onthe capacitors after the first charge sharing phase is as follows:

C₁₃ C₁₄ C₁₅ + C_(C) Charge (pico-Coulombs) 3.1034 2.8966 0 Voltage(Volts) 1.0345 1.0345 0

Referring to FIG. 9 c, the first mode of the second charge sharing phaseincludes opening the third and fourth switches 865, 866 and closing afifth switch 868, causing charge sharing between C₁₄ and the serialcombination of C_(C) and C₁₅. Upon completion of the second chargesharing phase, 0.141 pC of charge are transferred from C₁₄ to the serialcombination of C_(C) and C₁₅. The summary of charges and voltages on thecapacitors after the second charge sharing phase is as follows:

C₁₃ C₁₄ C₁₅ + C_(C) Charge (pico-Coulombs) 3.1034 2.756 0.141 Voltage(Volts) 1.0345 0.9842 0.9842

Referring to FIG. 9 d, the read phase connects a sixth switch 870 to aread line 872. Thus, the charge on C₁₄ is shared onto read line 872.Note that shared charge is proportional to the input voltage (i.e.,2.0V) times the filter coefficient (b₀=0.4593).

4.5.2 Second Charge Scaling Circuit

Referring to FIGS. 10 a-10 d, an example of using the second chargescaling circuit 233 of FIG. 3 to scale an input voltage of 2.0V by afilter coefficient b₁=0.9137 is illustrated. In the example presentedbelow, the configured capacitances of the capacitors shown in the figureare C₂₂=3.0 pF, C₂₄=0.2 pF, C_(C)=0.5 pF, and C₂₅=0.1 pF. As was thecase above, C_(C) is a capacitor which can be switched into or out ofthe charge scaling circuit 233 for the purpose of providing additionalscaling factors. The serial combination of C_(C) and C₂₅ has anequivalent capacitance of 0.083 pF.

Before the capacitor of the analog memory (i.e., C₂₂) is loaded, thecharges and voltages on all of the capacitors are assumed to be zero asis summarized in the following table:

C₂₂ C₂₄ C₂₅ + C_(C) Charge (pico-Coulombs) 0 0 0 Voltage (Volts) 0 0 0

Referring to FIG. 10 a, the capacitor in the first analog memory 230 ischarged in the memory loading phase. In this phase a first switch 962and a second switch 964 of the first analog memory 230 are closed,causing the input voltage of 2.0V to be applied to C₂₂. Applying 2.0V toC₂₂ results in a charge of 6.0 pC on C₂₂. Thus the summary of thecharges and voltages on the capacitors is as follows:

C₂₂ C₂₄ C₂₅ + C_(C) Charge (pico-Coulombs) 6.0 0 0 Voltage (Volts) 2.0 00

Referring to FIG. 10 b, the first charge sharing phase includes openingthe first and second switches 962, 964 and closing a third switch 965and a fourth switch 966, causing charge sharing between C₂₂ and C₂₄.Upon completion of the first charge sharing phase, 0.375 pC of chargeare transferred from C₂₃ to C₂₄. The summary of charges and voltages onthe capacitors after the first charge sharing phase is as follows:

C₂₂ C₂₄ C₂₅ + C_(C) Charge (pico-Coulombs) 5.625 0.375 0 Voltage (Volts)1.875 1.875 0

Referring to FIG. 10 c, the second mode of the second charge sharingphase includes leaving the third and fourth switches 965, 966 closed andclosing a fifth switch 968, causing charge sharing between C₂₂, C₂₄, andthe serial combination of C_(C) and C₂₅. Upon completion of the secondcharge sharing phase, 0.152 pC of charge are transferred from C₂₂ andC₂₄ to the serial combination of C_(C) and C₂₅. The summary of chargesand voltages on the capacitors after the second charge sharing phase isas follows:

C₂₂ C₂₄ C₂₅ + C_(C) Charge (pico-Coulombs) 5.482 0.365 0.152 Voltage(Volts) 1.8274 1.8274 1.8274

Referring to FIG. 10 d, the read phase connects a sixth switch 974 to aread line 972 while leaving the third, fourth, and fifth switches 965,966, 968 open.

Thus, the charge of 5.482 pC on C₂₂ is shared onto the read line 972.Note that the shared charge is proportional to the input voltage (i.e.,2.0V) times the filter coefficient (b₀=0.9137=5.482 pC/6.0 pC).

5 Alternatives

In the above examples, the loading and use of capacitors in the analogmemories 230, 240 is described as being carried out in a regularpattern. However, in some examples, to address defects or mismatches incapacitors, the loading and use of the capacitors can be carried out inan irregular or pseudo-random pattern.

In some examples, additional elements (e.g., switches) are added toaddress parasitic capacitances, including capacitances of switchelements (e.g., switching transistors) which can disrupt the applicationof filter coefficients to signal samples.

In some examples, a filter designer may specify a filter characteristicusing a computer program and then pass the specified filtercharacteristic to the filter configuration module which maps the filtercharacteristic to a configuration instruction which is usable by the IIRfilter modules.

While the examples above describe second order IIR filters, other filtertypes are possible. For example, some filtering systems include IIRfilters including only poles. In some examples, filtering systems mayinclude higher order filters and the degree of the numerator anddenominator polynomials are not required to be the same.

In some examples, the filter configuration module described abovecreates the configuration instructions/data by generating one or moreconfiguration words for each charge sharing circuit. For example, aconfiguration word for a particular charge sharing circuit may be a 14bit word which includes 6 bits for configuring a first capacitor, C₁, ofthe charge sharing circuit, 6 bits for configuring a second capacitor,C₂, of the charge sharing circuit, 1 bit for configuring the chargesharing mode of the charge sharing circuit, and 1 bit for configuringwhether the additional capacitor, C_(C), is coupled into the chargesharing circuit.

Note that with the switch configurations shown in FIG. 3, other modes ofoperation of the charge scaling circuit can be used, for example, withyet other combinations of switches being open and closed in two chargesharing phases, with more than two charge sharing phases, and/or withdifferent selections of capacitors for reading on output. Furthermore,it should be understood that the particular circuit arrangements shownfor the charge scaling circuits can be used, for example, with more thantwo configurable capacitors, which may also be arranged in a chain withintervening switches or provide full switchable connectivity between thecapacitors.

Note that in the description of the detailed operation of the example ofthe system as illustrated in FIG. 8, a single read phase is used totransfer all the charge in the selected capacitors to the capacitors atthe gain element. However, it should be using a single clock phase totransfer the charge, multiple clock phases can be used, and in each ofthese multiple phases different subsets of capacitors can transfer theircharge. In another alternative, the read phase can be divided into afirst phase in which all the capacitors are coupled so that they sharetheir charge, without coupling them to the amplifier, thereby resultingin a common voltage on all the capacitors. In a second phase of the readphase, it is not necessary to couple all the capacitors to theamplifier, and by selecting a subset of the capacitors, a selectablegain reduction can be achieve so that the effective gain is based notonly on the configuration of the capacitors at the amplifier, but alsothe selection of the capacitors for transfer of charge to the amplifiercapacitors.

In some examples, the gain of the amplifier described above can beoffset with the denominator coefficients. For example, all of thedenominator coefficients, a, can be scaled by 1/max(|a|) and the gain ofthe amplifier can be max(|a|), yielding the same filter characteristicthereby avoiding scaling change by factors that are greater than 1 or byvery small factors. Note an overall scaling of the transfer function canaddress the maximum magnitude of the numerator coefficients, which aretherefore all assume to be less than or equal to 1 in magnitude.

In some examples, the coefficients implemented by the charge sharingcircuits are represented digitally in the filter configuration system106, and are transformed to configure the charge sharing circuit toachieve a linear relationship between a desired coefficient and a degreeof charge transfer. For instance, a lookup table in the filterconfiguration system accepts a coefficient representation and providesoutputs that configure the charge sharing circuit.

The configuration instructions/data can be provided in a number ofdifferent ways. In some examples, each charge sharing circuit include avolatile digital storage register and/or a fixed data register (e.g.,metal layer ROM cells), as well as logic circuitry for combining theclock signal with stored values to control the switches. In someimplementations, when the system is powered up, the values form fixedregister are transferred to the volatile register, for example, toimplement a default filter. In other examples, on powerup, the systemretrieves the data to configure the circuits from an external memory,for example, over a serial connection. In some examples, the data valuesare set through control registers under external control.

In some examples, the filter configuration system 106 is implemented insoftware running on a workstation is used to determine configurationinstructions/data which are subsequently used to configure the filteringcircuits and systems described above. In some examples, theconfiguration instructions/data determined by the filter configurationsystem are transmitted directly to the filter system 100, while in otherexamples, the instructions/data (which essentially impart functionalityto the filtering circuits and systems described above) are stored on atangible media which is later used to transfer the configurationinstructions/data to the filter system.

It is to be understood that the foregoing description is intended toillustrate and not to limit the scope of the invention, which is definedby the scope of the appended claims. Other embodiments are within thescope of the following claims.

1. A signal processing device comprising a first discrete time analogsignal filter section, said first section comprising: an input foraccepting a time series of input signal values; an output for providinga time series of output signal values; an analog signal storage sectioncomprising a plurality of capacitors, the analog signal storage sectioncomprising a first storage section having a plurality of capacitorsconfigured for selective charging according to input signal values; afirst plurality of scaling sections, each scaling section beingassociated with a different part of the first storage section; switchingcircuit elements configurable to, for each time step of the time seriesof input signal values, (a) charge a subset of the capacitors of thefirst storage section according to the input signal value for that timestep, (b) in a series of one or more phases, couple multiple subsets oftwo or more capacitors each subset being selected from capacitors of oneof the scaling sections and capacitors in the part of the first storagesection associated with the one of the scaling section, and (c) couple agroup of capacitors to form an intermediate signal value, the group ofcapacitors including at least one capacitors from the analog signalstorage section and at least one capacitor from the first plurality ofscaling sections, circuitry for forming each output signal value of thetime series of output signal values from a corresponding intermediatesignal value; control logic for controlling configuration of theswitching circuit elements in successive phases of a clock signal toform the time series of output signal values as an application of afilter to the time series of input signal values.
 2. The device of claim1 wherein the signal storage section further comprises a second storagesection having a plurality of capacitors configured for selectingcharging according to intermediate signal values and a second pluralityof scaling sections, each scaling section being associated with adifferent part of the second storage section, wherein the switchingcircuit elements are further configurable to, for each formedintermediate signal value, (a) charge a subset of capacitors of thesecond storage section according to the intermediate signal value, (b)in a series of one or more phases, couple multiple subsets of two ormore capacitors, each subset being selected from capacitors of one ofthe scaling sections of the second plurality of scaling sections andcapacitors in the part of the second storage section associated with theone of the scaling section, and wherein the groups of capacitors coupledto form the intermediate signal values includes at least one capacitorfrom the second storage section and at least one capacitor from thesecond plurality of scaling sections.
 3. The device of claim 1 furthercomprising active circuitry for charging the subsets of subsets ofcapacitors of the second storage section according to the intermediatesignal values. 4.-10. (canceled)
 11. A method for operating a discretetime analog signal filter section of an integrated circuit in successivephases of a clock signal, the method comprising, for each time step of aseries of input signal values: accepting an input signal value; in afirst phase, configuring switching circuit elements to charge a subsetof capacitors of an analog signal storage section, including charging asubset of capacitors of a first storage section of the analog signalstorage section according to the input signal value; in a second phase,configuring the switching circuit elements to couple multiple subsets oftwo or more capacitors each subset being selected from capacitors of onescaling section of a first plurality of scaling sections and capacitorsof a part of the first storage section associated with the one of thescaling section, and in a third phase, configuring the switching circuitelements to couple a group of capacitors to form an intermediate signalvalue, the group of capacitors including at least one capacitor from theanalog signal storage section and at least one capacitor from the firstplurality of scaling sections; forming an output signal value accordingto the intermediate value; wherein configuring the switching circuitelements includes controlling configuration of the switching circuitelements in successive time steps to form the time series of outputsignal values as an application of a filter to the time series of inputsignal values.
 12. The method of claim 11 further comprising, for eachformed intermediate signal value, configuring the switching circuitelements to charge a subset of capacitors of a second storage section ofthe analog signal storage section according to the intermediate signalvalue, and in a series of one or more phases, couple multiple subsets oftwo or more capacitors-each subset being selected from capacitors of oneof the scaling sections of a second plurality of scaling sections andcapacitors of a part of the second storage section associated with theone of the scaling section, and wherein the groups of capacitors coupledto form the intermediate signal values includes at least one capacitorfrom the second storage section and at least one capacitor from thesecond plurality of scaling sections. 13-19. (canceled)
 20. The signalprocessing device of claim 1 wherein the control logic is configured toform the time series of output signal values as an application of afinite-impulse-response (FIR) filter to the time series of input values.21. The signal processing device of claim 2 wherein the control logic isconfigured to form the time series of output signal values as anapplication of an infinite-impulse-response (IIR) filter to the timeseries of input values.
 22. The method of claim 11 wherein forming thetime series of output signal values comprises forming said time seriesas an application of a finite-impulse-response (FIR) filter to the timeseries of input values.
 23. The method of claim 12 wherein forming thetime series of output signal values comprises forming said time seriesas an application of an infinite-impulse-response (IIR) filter to thetime series of input values.